AD834

REV. C

–5–

BASIC OPERATION

Figure 7 is a functional equivalent of the AD834. There are

three differential signal interfaces: the voltage inputs X =

X1–X2 and Y = Y1–Y2, and the current output, W (see Figure

7) which flows in the direction shown when X and Y are posi-

tive. The outputs W1 and W2 each have a standing current of

typically 8.5 mA.

Figure 7. AD834 Functional Block Diagram

The input voltages are first converted to differential currents

which drive the translinear core. The equivalent resistance of

the voltage-to-current (V-I) converters is about 285

Ω. This low

value results in low input related noise and drift. However, the

low full-scale input voltage results in relatively high nonlinearity

in the V-I converters. This is significantly reduced by the use of

distortion cancellation circuits which operate by Kelvin sensing

the voltages generated in the core—an important feature of the

AD834.

The current mode output of the core is amplified by a special

cascode stage which provides a current gain of nominally

× 1.6,

trimmed during manufacture to set up the full-scale output cur-

rent of

±4 mA. This output appears at a pair of open

collectors which must be supplied with a voltage slightly

above the voltage on Pin 6. As shown in Figure 8, this can be

arranged by inserting a resistor in series with the supply to this

pin and taking the load resistors to the full supply. With R3 =

60

Ω, the voltage drop across it is about 600 mV. Using two

50

Ω load resistors, the full-scale differential output voltage is

±400 mV.

The full bandwidth potential of the AD834 can only be realized

when very careful attention is paid to grounding and decou-

pling. The device must be mounted close to a high quality

ground plane and all lead lengths must be extremely short, in

keeping with UHF circuit layout practice. In fact, the AD834

shows useful response to well beyond 1 GHz, and the actual up-

per frequency in a typical application will usually be determined

by the care with which the layout is effected. Note that R4 (in

duces a voltage drop of about 150 mV. It is made large enough

to reduce the Q of the resonant circuit formed by the supply

lead and the decoupling capacitor. Slightly larger values can be

used, particularly when using higher supply voltages. Alterna-

tively, lossy RF chokes or ferrite beads on the supply leads may

be used.

Figure 8 shows the use of optional termination resistors at the

inputs. Note that although the resistive component of the input

Figure 8. Basic Connections for Wideband Operation

impedance is quite high (about 25 k

Ω), the input bias current of

typically 45

µA can generate significant offset voltages if not

compensated. For example, with a source and termination

resistance of 50

Ω (net source of 25 Ω) the offset would be

25

Ω × 45 µA = 1.125 mV. This can be almost fully cancelled by

including (in this example) another 25

Ω resistor in series with

the “unused” input (in Figure 8, either X1 or Y2). In order to

minimize crosstalk the input pins closest to the output (X1 and

Y2) should be grounded; the effect is merely to reverse the

phase of the X input and thus alter the polarity of the output.

TRANSFER FUNCTION

The output current W is the linear product of input voltages

X and Y divided by (1 V)

current” of 4 mA:

W

=

XY

1V

4 mA

Provided that it is understood that the inputs are specified in

volts, a simplified expression can be used:

W

=(XY )4 mA

Alternatively, the full transfer function can be written:

W

=

XY

1V

×

1

250

Ω

When both inputs are driven to their clipping level of about

1.3 V, the peak output current is roughly doubled, to

±8 mA,

but distortion levels will then be very high.

TRANSFORMER COUPLING

In many high frequency applications where baseband operation

is not required at either inputs or output, transformer coupling

can be used. Figure 9 shows the use of a center-tapped output

transformer, which provides the necessary dc load condition at

the outputs W1 and W2, and is designed to match into the de-

sired load impedance by appropriate choice of turns ratio. The

specific choice of the transformer design will depend entirely on

the application. Transformers may also be used at the inputs.

Center-tapped transformers can reduce high frequency distor-

tion and lower HF feedthrough by driving the inputs with

balanced signals.