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MC33067 Scheda tecnica(PDF) 10 Page - ON Semiconductor |
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MC33067 Scheda tecnica(HTML) 10 Page - ON Semiconductor |
10 / 16 page MC34067 MC33067 10 MOTOROLA ANALOG IC DEVICE DATA APPLICATIONS INFORMATION The MC34067 is specifically designed for zero voltage switching (ZVS) quasi–resonant converter (QRC) applications. The IC is optimized for double–ended push–pull or bridge type converters operating in continuous conduction mode. Operation of this type of ZVS with resonant properties is similar to standard push–pull or bridge circuits in that the energy is transferred during the transistor on–time. The difference is that a series resonant tank is usually introduced to shape the voltage across the power transistor prior to turn–on. The resonant tank in this topology is not used to deliver energy to the output as is the case with zero current switch topologies. When the power transistor is enabled the voltage across it should already be zero, yielding minimal switching loss. Figure 19 shows a timing diagram for a half–bridge ZVS QRC. An application circuit is shown in Figure 20. The circuit built is a dc to dc half–bridge converter delivering 75 W to the output from a 48 V source. When building a zero voltage switch (ZVS) circuit, the objective is to waveshape the power transistor’s voltage waveform so that the voltage across the transistor is zero when the device is turned on. The purpose of the control IC is to allow a resonant tank to waveshape the voltage across the power transistor while still maintaining regulation. This is accomplished by maintaining a fixed deadtime and by varying the frequency; thus the effective duty cycle is changed. Primary side resonance can be used with ZVS circuits. In the application circuit, the elements that make the resonant tank are the primary leakage inductance of the transformer (LL) and the average output capacitance (COSS) of a power MOSFET (CR). The desired resonant frequency for the application circuit is calculated by Equation 6: L L 2CR (6) 1 = π 2 ƒr In the application circuit, the operating voltage is low and the value of COSS versus Drain Voltage is known. Because the COSS of a MOSFET changes with drain voltage, the value of the CR is approximated as the average COSS of the MOSFET. For the application circuit the average COSS can be calculated by Equation 7: measured at (7) CR 1 2 2 * COSS = in V The MOSFET chosen fixes CR and that LL is adjusted to achieve the desired resonant frequency. However, the desired resonant frequency is less critical than the leakage inductance. Figure 19 shows the primary current ramping toward its peak value during the resonant transition. During this time, there is circulating current flowing through the secondary inductance, which effectively makes the primary inductance appear shorted. Therefore, the current through the primary will ramp to its peak value at a rate controlled by the leakage inductance and the applied voltage. Energy is not transferred to the secondary during this stage, because the primary current has not overcome the circulating current in the secondary. The larger the leakage inductance, the longer it takes for the primary current to slew. The practical effect of this is to lower the duty cycle, thus reducing the operating range. The maximum duty cycle is controlled by the leakage inductance, not by the MC34067. The One–Shot in the MC34067 only assures that the power switch is turned on under a zero voltage condition. Adjust the one–shot period so that the output switch is activated while the primary current is slewing but before the current changes polarity. The resonant stage should then be designed to be as long as the time for the primary current to go to zero amps. |
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